Efficient rectifier circuit operating at N78 and N79 sub-6 GHz 5G bands for microwave energy-harvesting and power transfer applications

Abstract

The microwave energy-harvesting (MEH) and microwave power transfer (MPT) technologies have become the most emerging areas of research nowadays. The microwave rectifier circuit is the bottleneck of both the MEH and MPT systems. The efficiency of the system depends on the power conversion efficiency (PCE) of the rectifier. Due to the recent advancement of the fifth-generation communication system, it is desirable to propose an efficient rectifier operating at sub-6 GHz 5G bands. A dual-band rectifier circuit is designed and demonstrated for MEH/MPT purposes, specifically at sub-6 GHz 5G frequency bands. The dual-band matching is achieved by using a stepped impedance transmission line. The rectifier covers N78 (3.3–3.6 GHz) and N79 (4.8–5.0 GHz) bands. Peak PCE of 67.6% @ 3.5 GHz and 56.8% @ 4.9 GHz are achieved. For validation purpose, the rectifier is fabricated and characterized and measured results show good agreement with simulated results.

Introduction

With the invention of rectenna (rectifying antenna), microwave energy-harvesting (MEH) and microwave power transfer (MPT) technologies have received wide notice in the research and industry. Rectennas give MEH and MPT systems flexibility in terms of physical wiring, battery charging, and periodic replacement. Such systems have found applications in wireless charging of mobile phones or any electric/electronic equipment, radio-frequency (RF) identification (RFID) tags, Internet of Things, implantable/wearable biomedical devices, wireless sensors, and devices in the smart home [1–3]. Wi-Fi, GSM, LTE, and third/fourth/fifth generation (3G/4G/5G) are some wireless communication systems that radiate electromagnetic power in the environment. Radiation from these systems may serve as a source of electromagnetic energy that may harvest using an appropriate rectenna. Rectifying antenna A rectenna consists of an antenna, and rectifier, through a matching circuit. The rectifier is the crucial component of the rectenna for converting microwave energy into electrical direct current (DC). Therefore, several recent studies have been conducted to introduce efficient rectifier circuits [4–8]. The popular rectifier topologies are series diode [9], shunt diode [10], voltage doubler (VD) [11], and Greinacher full-wave rectifiers [12]. VD topology doubles the output voltage and improves the power handling capability for low power requirements [13]. The matching circuit is very essential for maximum power transfer from the antenna to the rectifier. To know the input impedance of the rectifier is necessary for proper matching network (MN) design. The operating frequency and input power level affect the input impedance of the rectifier. Hence, it is very difficult to find impedance matching at two or more frequencies simultaneously [14–16]. A dual-band rectifier (DBR) has the potential to utilize two bands simultaneously which can improve the output power levels. In recent years, many techniques have been adopted to design a DBR. In paper [17], a rectifier using an impedance matching network (IMN), having three lumped inductors and a radial stub, was designed to work at 0.915 and 2.45 GHz, simultaneously. A dual-band matching was realized using cascaded L-type and π-type networks to design a DBR circuit working at 0.915 and 2.45 GHz bands [18]. In paper [19], two-branch IMN was utilized to match within the 0.9 GHz UHF and 2.45 GHz ISM bands using a T-type MN. In paper [20], a DBR circuit consisting of a band-stop filter (BSF), a short-ended quarter-wave transmission line (QWTL) in series with a shunt diode, and a short-ended quarter-wave stub was proposed. Here, BSF and QWTL were applied to block the second harmonic at 2.45 and 5.8 GHz, respectively. A short-ended quarter-wave stub was applied for impedance tuning at the first band. In papers [21, 22], dual-band IMN at 2.45 and 5.8 GHz has been designed by tuning the characteristic impedance (width) of a quarter-wave short-ended stub. A dual-band matching at 0.915 and 2.45 GHz was realized using a stepped impedance stub (SIS) [23]. The uniform width stub was cut out to form a SIS to find matching at two resonance bands to design a DBR. In paper [24], a DBR was designed using lumped component-based modified T-section IMN. The DBRs above targeted GSM 0.9 GHz, GSM 1.8 GHz, ISM 2.45 GHz, and ISM 5.8 GHz bands. However, the commercialization of fifth-generation (5G) wireless communication has gained attention to the design of a rectifier operating at sub-6 GHz 5G bands. Some of the DBRs are reported at one of the 5G bands. DBR was reported to operate at 2.45 GHz Wi-Fi and 3.5 GHz 5G using composite left/right-handed-based IMN [25], short-ended stub IMN [26], and two-branch IMN [27]. In [28–30], DBRs were proposed to work at 3.5 and 5.8 GHz bands. In paper [31], a rectifier has been designed using π-type dual-band IMN with a dual-band harmonic termination network (HTN). In paper [32], a DBR using two-branch IMN was utilized to work at 2.45 and 5 GHz. In paper [33], a three-stage Villard DBR was designed to operate at 2.4 and 5.2 GHz.

All of these rectifiers have relatively low efficiencies. Also, the DBR reported in papers [25–33] does not focus on dual 5G bands. In paper [34], a Greinacher full-wave rectifier circuit was developed to operate at dual 5G bands, and at 12 dBm input, it achieved 22.7% and 30.7% power conversion efficiency (PCE) at 3.5 and 5.0 GHz, respectively. In paper [35], a transparent and flexible dual-band 5G rectifier was reported, but the achieved PCE at 13 dBm was 54.67% and 7% at N78 and N79 bands, respectively, which is very poor, especially at the N79 band. Hence, it is desirable to develop an efficient and compact rectifier circuit to harvest RF energy of sub-6 GHz dual 5G bands. The 5G bands adopted in some nations are displayed in Table 1 [36] which is self-explanatory.

Table 1. Sub-6 GHz fifth-generation bands

The short-ended stub [20–22], stepped impedance short-ended stub [23], composite right/left-handed structure [25], half-wavelength transmission line (TL) [30], π-type dual-band IMN with dual-band HTN [31], cross-branch dual-stub [34], and combination of open and short-ended stub [37] were utilized in some of the existing designs to achieve dual-band matching. The proposed design uses a novel stepped impedance TL technique to satisfy the matching at two sub-6 GHz fifth-generation (5G) bands. Hence, the novelty of the proposed dual-band 5G rectifier is in terms of (a) dual 5G bands with better performance compared to existing ones and (b) a novel dual-band matching technique using a stepped impedance TL.

This paper demonstrates an efficient DBR circuit covering 3.3–3.7 GHz (N 78 band adopted by India, China, Europe, and Korea) and 4.8–5.0 GHz bands (N 79 band adopted by China). The power/frequency-weighted efficiency (PFE) has been derived to compare the overall circuit performance including frequency of operation and operating power level. The proposed work exhibits the highest PFE. The rest of this paper is arranged as follows. The “Dual-band rectifier” discusses the geometric description and simulated results of the proposed DBR circuit. The “Fabrication, measurement, and comparison” section discusses the fabrication and characterization of the rectifier. Finally, the work is concluded in the “Conclusion” section.

Dual-band rectifier

The design of a DBR is auspicious to operate at sub-6 GHz 5G bands. For this, a dual-band MN is needed. It is very challenging to design an MN satisfying two different operating frequencies because of the diode’s nonlinear nature. One of the methods is to combine two rectifiers operating at two different operating frequencies. The second method is to utilize a single dual-band MN. The proposed design uses a novel stepped impedance TL to satisfy the matching at two sub-6 GHz fifth-generation (5G) bands.

Geometrical description

Figure 1(a) and (b) displays the schematic and layout view of the proposed DBR circuit. The SOT-23 package of HSMS-2862 (VD topology) is used in the proposed rectifier. The substrate used in this design is RO4003 (dielectric constant = 3.55, loss tangent = 0.0027, and thickness = 1.524 mm). The MN is based on stepped impedance TL segments of different lengths. For simplification, the principle of dual-band impedance transformation is analyzed using lossless TL theory [38]. The impedance ${Z_{\textrm{a}}}$ across the TL, TL₁ can be calculated using (1):

(1)\begin{equation}{Z_{{\textrm{a}}@{f_i}}} = {Z_{0{\textrm{TL}}1}}{{{Z_{{\textrm{L}}@fi}} + j{Z_{0{\textrm{TL}}1}}\tan {\beta _{fi}}{L_{{\textrm{TL}}1}}} \over {{Z_{0{\textrm{TL}}1}} + j{Z_{{\textrm{L}}@fi}}\tan {\beta _{fi}}{L_{{\textrm{TL}}1}}}}\end{equation}

Figure 1. Image of dual-band rectifier: (a) Schematic view and (b) Layout view.

Here, “i” denotes 1 and 2 of the DBR’s first and second bands, f ₁ and f ₂, respectively. The impedance ${Z_{\textrm{b}}}$, ${Z_{\textrm{c}}}$, and ${Z_{{\textrm{in}}}}$ across the TL₂, TL₃, and TL₄, respectively, can be calculated using (2), (3), and (4), respectively:

(2)\begin{equation}{Z_{{\textrm{b}}@{f_i}}} = {Z_{0{\textrm{TL}}2}}{{{Z_{{\textrm{a}}@fi}} + j{Z_{0{\textrm{TL}}2}}\tan {\beta _{fi}}{L_{{\textrm{TL}}2}}} \over {{Z_{0{\textrm{TL}}2}} + j{Z_{{\textrm{a}}@fi}}\tan {\beta _{fi}}{L_{{\textrm{TL2}}}}}}\end{equation}

(3)\begin{equation}{Z_{{\textrm{c}}@{f_i}}} = {Z_{0{\textrm{TL}}3}}{{{Z_{{\textrm{b}}@fi}} + j{Z_{0{\textrm{TL}}3}}\tan {\beta _{fi}}{L_{{\textrm{TL}}3}}} \over {{Z_{0{\textrm{TL}}3}} + j{Z_{{\textrm{b}}@fi}}\tan {\beta _{fi}}{L_{{\textrm{TL3}}}}}}\end{equation}

(4)\begin{equation}Z_{in@f_i}=Z_{0TL4}\frac{Z_{c@fi}+jZ_{0TL4}\tan\beta_{fi}L_{TL4}}{Z_{0TL4}+jZ_{c@fi}\tan\beta_{fi}L_{TL4}}\end{equation}

For matching at two frequencies, f ₁ and f ₂, the real and imaginary parts of the input impedance ${Z_{{\textrm{in}}}}$should follow the criteria of (5) and (6).

(5)\begin{equation}{{\rm Re}}\left( {{Z_{{{\rm in}}@{f_1}}}} \right) \approx {{\rm Re}}\left( {{Z_{{{\rm in}}@{f_2}}}} \right) \approx 50\;\Omega \end{equation}

(6)\begin{equation}{{\rm Im}}\left( {{Z_{{{\rm in}}@{f_1}}}} \right) \approx {{\rm Im}}\left( {{Z_{{{\rm in}}@{f_2}}}} \right) \approx 0\;\Omega \end{equation}

The matching at two frequency bands is achieved by tuning the TL parameters. The 3.5 and 4.9 GHz dual sub-6 GHz 5G bands are selected as f ₁ and f ₂, respectively.

Figure 2(a) shows the input impedance of the rectifier circuit without using the MN are Z _L@f1 = 157 + j8 Ω and Z _L@f2 = 20 – j61 Ω. After introducing a MN using the source pulling technique, Fig. 2(b) shows the input impedance of the rectifier becomes, Z _{[email protected] GHz} = 52-j1 Ω and Z _{[email protected] GHz} = 45 – j9 Ω which satisfies (5) and (6). The optimal dimensions of the TLs (in mm) are; L _TL1 = 4.5, L _TL2 = 4.2, L _TL3 = 5, L _TL4 = 3.8, L _TL5 = 6, L _TL6 = 6.4, L _TL7 = 3.8, W _TL1 = 1, W _TL2 = 8.6, W _TL3 = 7, W _TL4 = 3.6, W _TL5 = W _TL6 = 1, W _TL7 = 1.1, radius of fan-shaped stub r = 9 and angle θ = 60^o.

Figure 2. Input impedance of rectifier (a) without MN (Z _L) and (b) with an MN (Z _in).

Simulated results and discussion

The rectifier circuit is simulated using Keysight advanced design system. At different values of input power levels, the simulated S ₁₁ vs. frequency is depicted in Fig. 3(a). The result shows that the circuit is well-matched at both 3.5 GHz and 4.9 GHz 5G bands. The PCE in the desired band is simulated for different load values and input power. The PCE vs. P _in at 3.5 and 4.9 GHz is displayed in Fig. 3(b). The maximum PCE is 70.6% @6 dBm and 62.9% @ 5 dBm at 3.5 and 4.9 GHz, respectively, for a 3 kΩ load. The simulated PCE is greater than 50% for the input power range from −2.5 to 12.4 and −0.8 to 8.3 dBm for 3.5 and 4.9 GHz, respectively. PCE vs. load is depicted in Fig. 3(c) at 0, 3, and 6 dBm input power levels at both frequencies. With the increase of load value, PCE starts increasing and reached its maximum value, and then starts decreasing. Hence, a 3-kΩ resistor is selected as the optimum load to better perform at an input power level between −10 and 10 dBm.

Figure 3. Simulated performance: (a) S ₁₁ vs. frequency, (b) PCE vs. input power (P _in), (c) PCE vs. load (R _L). (d) PCE vs. frequency, and (e) V _out vs. frequency.

Figure 4. (a) Measurement setup, (b) rectifier prototype.

Figure 3(d) shows the PCE vs. frequency graph. The maximum simulated PCE at two different bands is 72.8% and 67.2% at 3.55 and 4.94 GHz at 6 dBm. PCE is higher than 60%, from 3.35 to 3.83 and 4.83 to 5.09 GHz. At a 3 dBm power level, the PCE is higher than 55% from 3.3 to 3.74 and 4.8 to 5.0 GHz. PCE is 61.4% and 53.5% at 3.45 and 4.8 GHz, respectively, at 0 dBm. The V _out vs. frequency at −3, 0, 3, and 6 dBm input power levels are displayed in Fig. 3(e). At 3 dBm, the V _out is greater than 1.5 V for both bands from 3.1 to 3.84 and 4.6 to 5.07 GHz. Also, the V _out is more than 2.5 V from 3.28 to 3.86 and 4.76 to 5.13 GHz.

Fabrication, measurement, and comparison

For the endorsement of the proposed 5G rectifier, an experimental setup is depicted in Fig. 4(a). Figure 4(b) describes the fabricated prototype of the proposed DBR. The RF signal generator was used for input power feeding to the rectifier circuit, and a digital meter was used for output voltage measurement. The output DC voltage was noted, and the PCE was calculated using (7).

(7)\begin{equation}{\textrm{PCE}} = {{V_{{\textrm{out}}}^2} \over {{R_{\textrm{L}}}*{P_{{\textrm{in}}}}}} \times 100\% \end{equation}

Figure 5(a) shows the measured S11 (bottom left negative y-axis), Vout (right y-axis), and PCE (top left positive y-axis) vs. frequency, respectively at 6 dBm input power. Load resistance of 3 kΩ and input power of 0 dBm are applied to the rectifier. The measured value of |S ₁₁| is −28.9 and −16.2 dB at 3.7 and 5.0 GHz respectively. The maximum recorded DC output voltage is 2.96 and 2.77 V at 3.7 and 5.05 GHz, respectively. The PCE is more than 50% for 3.25–3.96 and 4.81−5.15 GHz bands. Figure 5(b) and (c) show the V _out and PCE vs. P _in, respectively. The measured DC output voltages recorded are 1.32 V @ 0 dBm and 2.53 V @ 5 dBm at 3.5 GHz, respectively. At 4.9 GHz, the output voltages are 1.26 V @ 0 dBm and 2.31 V @ 5 dBm. At 3.5 GHz, the measured PCE is 58.1% @ 0 dBm, 67.3% @ 5 dBm, and the maximum PCE reached 67.6% @ 6 dBm input power. The measured value of PCE at 4.9 GHz is 52.8% at 0 dBm, 56.3% at 5 dBm, and the peak PCE reached 56.8% at 3.5 dBm input power. Measured results show good agreement with the simulated ones. The little difference between simulated and measured performance may be due to diode model inaccuracies and fabrication tolerance.

Figure 5. Measured results of (a) S ₁₁, PCE, V _out vs. frequency, (b) V _out, vs. P _in, (c) PCE vs. P _in.

Table 2 describes the comparison of the 5G DBR performance with some existing works. The rectifier achieved 58.1% and 52.8% PCE at 0 dBm for operating frequencies 3.5 and 4.9 GHz, respectively. The PCE at 0 dBm of existing rectifiers presented in papers [24, 26, 27, 39] looks better because of the lower operating frequency (band). Because of the skin effect and parasitic effects, the loss increases at a high operating frequency. Therefore, to compare DBRs operating at different bands, a frequency-weighted efficiency (FE) can be calculated using (8) [39]. The PCE presented in papers [31, 39] looks better because of the higher input power level. As the PCE of the rectifier circuit is also influenced by input power level, a new PFE is presented in (9) to compare the overall circuit performance.

(8)\begin{equation}{{\rm FE}} = \sum\limits_{i = 1}^n {\left[ {{\eta _i}\,f_i^{0.25}} \right]} \end{equation}

(9)\begin{equation}{\textrm{PFE}} = {1 \over {{P_{{\textrm{in}}}}^{0.1}}} \times {\textrm{FE}}\end{equation}

Table 2. Comparison with existing works

FE: frequency-weighted efficiency FE. PFE: power/frequency-weighted efficiency.

where n is the number of bands, ${f_i}$ is the ith frequency band, ${\eta _i}$ is the PCE at ith frequency band, and ${P_{{\textrm{in}}}}$ is input power level in mW.

The power factor of input power in the denominator of (9) is decided based on the proposed circuit FOM at two different power levels (0 and 5 dBm). The circuit performance is found approximately equal using this overall PFE (i.e., approximately 158 and 156 at 0 and 5 dBm, respectively).

Conclusion

A rectifier to operate at dual sub-6 GHz 5G band has been designed. An HSMS2862 in the SOT-23 package (VD configuration) has been employed for RF to DC conversion. A stepped impedance TL-based MN has been utilized to transform the rectifier input impedance close to 50 Ω at both operating bands. The rectifier efficiently covers the 3.5 GHz (N78 band) and 4.9 GHz (N79 band) bands of fifth-generation wireless communication. A maximum PCE of 67.6% at 3.5 GHz and 56.8% at 4.9 GHz has been achieved. These characteristics make this rectifier a good candidate for MEH and MPT systems. Future work includes improving the sensitivity of the rectifier and a dual-band 5G antenna design to implement a rectenna circuit.